How To Generate Square Wave In Matlab

68892
Fırçasız doğru akım motorlu tahrik sistemlerinde oniki darbeli sürücü / Twelve-step drive of brushless dc machines
Yazar:LATİF TEZDUYAR
Danışman: PROF. DR. M. EMİN TACER
Yer Bilgisi: İstanbul Teknik Üniversitesi / Fen Bilimleri Enstitüsü / Elektrik Mühendisliği Ana Bilim Dalı
Konu:Elektrik ve Elektronik Mühendisliği = Electrical and Electronics Engineering
Dizin:Doğru akım motorları = Direct current motors ; Momentum = Momentum ; Sürücüler = Drivers ; Tahrik sistemleri = Propulsion systems Onaylandı
Doktora
Türkçe
1997
281 s.

ÖZET Bu tez çalışmasının amacı, sürekli mıknatıslı fırçasız motorlu tahrik sistemlerinde moment darbesi bileşeninin en azlanmasına, başka bir ifadeyle moment kalitesinin yükseltilmesine yönelik farklı bir besleme ve denetim yaklaşımı geliştirmektir. Literatürde verilen çalışmalardan ve ticari uygulamaya dönük uygulamalardan, sürekli mıknatıslı fırçasız motorların beslenmesinde temel olarak üç teknikten yararlanıldığı gözlenmektedir. Söz konusu besleme teknikleri sırasıyla, sinusoidal ve 120°- 180° elektriksel iletimli altı darbeli trapezoidal yaklaşımlar olarak tanımlanırlar. Bu bağlamda sürekli mıknatıslı fırçasız motorlu tahrik sistemlerini, besleme yapısına göre sinusoidal ve trapezoidal olarak ikiye ayırmak mümkündür. Tez çalışmasında, sürekli mıknatıslı fırçasız tahrik sistemi uygulaması, özel bir uygulamaya yönelik dış rotorlu doğrudan tahrikli bir motor içermektedir. Yukarıda anılan besleme seçeneklerinin tümü analitik veya durum uzayı yaklaşımını temel alan genelleştirilmiş matematik modeller yardımıyla analiz edilmiş ve sonuçlar irdelenmiştir. Analitik yöntemlerle yapılan irdelemeler sonucunda, sinusoidal yaklaşımın yüksek maliyetli ve karmaşık denetim yapısı gerektirdiği anlaşılmıştır. Bundan sonraki aşamada, altı darbeli trapezoidal 120° ve 180° elektriksel iletimli besleme tekniklerinin durum uzayı yaklaşımı temelli matematik modelleri kurulmuş ve sayısal benzetim sonuçlan analiz edilmiştir. İki besleme yaklaşımının sonuçlarının özel uygulama ölçeğinde irdelenip, yorumlanmasını takiben moment darbesi bileşeninin en azlanmasına yönelik farklı bir besleme tekniği geliştirilmiş ve bu tekniğe oniki darbeli besleme yaklaşımı adı verilmiştir. Oniki darbeli besleme yaklaşımının, bir sonraki aşamada matematik modeli kurulmuş ve model dördüncü dereceden Runge-Kutta algoritması yardımıyla çözülmüştür. Sayısal benzetim sonuçlarından, yeni besleme yaklaşımının uygulanması ile moment darbesi bileşen genliğinin, altı darbeli besleme tekniğine kıyasla yarıya indirildiği saptanmıştır. Bu bağlamda oniki darbeli besleme yaklaşımının pratik uygulamasında akım denetimini sağlayacak ve bu besleme şekline özel, histeresis temelli sabit frekanslı ve doğru akım harasında bir ve yalnız bir algılayıcı içeren, bir akım denetim tekniği geliştirilmiştir. Oniki darbeli besleme ve ona özel akım denetim yaklaşımının Matlab ortamında modelleri kurulmuş ve sayısal benzetim sonuçlan irdelenmiştir. Matlab sonuçlarından, doğru akım harasında yer alan tek algılayıcı ile akım denetiminin başarıldığı anlaşılmaktadır. Ayrıca geliştirilen akım denetimi altı darbeli 120° elektriksel iletimli besleme yaklaşımına da uygulanmış ve bu besleme seçeneğine de katkıları ortaya konulmuştur. Tez çalışmasında yukarıda verilen teorik irdelemenin ardından, altı ve oniki darbeli besleme ve geliştirilen akım denetim yaklaşımı, prototip olarak gerçekleştirilmiş ve tahrik sisteminin tüm hız aralığında temel deneyleri tamamlanmıştır. Böylelikle, ortalama moment anlamında sayısal benzetim ve deney sonuçlarının karşılaştırılması ve doğrulanması amaçlanmıştır. Tez çalışmasında özetle, oniki darbeli besleme ve ona özel geliştirilen sayısal temelli akım denetim tekniğinin uygulanması sonucunda, sürekli mıknatıslı fırçasız motorlu tahrik sistemlerinde moment darbesinin önemli oranda azaltılması ve doğru akım harasında bir ve yalnız bir algılayıcı üzerinden akım denetiminin sağlanması başarılmıştır.

The objective of this thesis which is called "Twelve-Step Drive of Brushless DC Machines" is to develop a new and general drive technique including commutation and current control for permanent magnet brushless machines to minimize the effects of electronic commutation and torque ripple which is the extension of discrete exication due to commutation. Advances in magnetic materials and improvements in silicon technology give new opportunities for permanent magnet brushless motors to move into new application areas. These improvements can be extended by more precise control strategies to improve the performance of these motors. As a result, nowadays there is an increasing tendency to make use of permanent magnet brushless machines in the control of variable-speed high performance applications where torque smoothness is essential. For example, the quality of the surface finish achievable with metal- working machine tools is directly dependent on the smoothness of the instantaneous torque delivered to the rotary tool-piece. In similar manner, the performance specifications of servo motors, which are used in equipment ranging from robots to satellite trackers require minimization of all sources of pulsating torque or torque ripple. Even mass-produced consumer products such as white goods or traction drives demand high levels of ripple free torque and low noise levels to meet user expectations. However, the cost of permanent magnet material will probably always limit the universal use of these motors. As power ratings increase, there comes a point where it is more cost effective to use induction motors or switched reluctance motors. However, this is not a hard task and fast limit as very effective brushless permanent magnet motors can be designed for high power ranges. Permanent magnet brushless motors are candidates for many high-performance applications such as those identified above because of their attractive characteristics in such key categories as power density, torque to inertia and current ratio, noise level, and electrical efficiency. There are two major classes of permanent magnet brushless motor (PMBM) drives which can be characterised by the shapes of their respective back-EMF waveforms that can be defined as sinusoidal and trapezoidal drives respectively. Under idealised conditions, each of these two types of PMBM drives is capable of producing perfectly smooth instantaneous torque waveforms. All is required to understand the torque production of PMBM drives, is the basic knowledge of electronic commutation and the working principle of step motors. This simple operating philosophy makes PMBM drives attractive for many high- performance applications. Although the operating principle seems very simple, the torque production mechanism due to electronic commutation or discrete excitationsteps have many specific and scientific problems to be solved. In this manner, only a few of the key relevant characteristics of these sinusoidal and trapezoidal PMBM drives will be briefly reviewed in the following paragraphs. Sinusoidal PMBM drives share many of the basic characteristics of other classic types of polyphase ac machine drive systems. Basically, both the machine back-EMF and current excitation waveforms are perfectly sinusoidal for ideally smooth torque generation. Sinusoidal back-EMF waveforms require that motor's stator windings be sinusoidally distributed around the airgap and/or the radial magnetic flux density amplitude generated by the rotor permanent magnets varies sinusoidally around the airgap. Rotors of sinusoidal permanent magnet brushless motors can be designed either surface-mounted or interior magnet configurations. Sinusoidal phase currents are typically developed using a current-regulated inverter that requires individual phase current sensors and a high-resolution rotor position sensor like resolver or precise encoder to maintain accurate synchronisation of the excitation waveforms with the rotor angular position at any special time instant. Any source of non-ideal properties which causes either the phase current or the back-EMF waveforms to change from their purely sinusoidal shapes, will typically be a reason to the production of undesired pulsating torque components. Trapezoidal PMBM drives, also known as brushless dc or electronically commutated motor drives, have some major differences reference to their sinusoidal counterparts. These machines are designed to develop trapezoidal back-EMF waveforms. It is common to enlarge the flat portion of the trapezoidal back-EMF waveform in trapezoidal PMBM drives to meet the ideal conditions, which are given in textbooks for smooth torque generation [1-5]. To meet this requirement, there is a general tendency in designing the trapezoidal motor with surface-mounted magnets and concentrated stator windings in contrast to the distributed windings preferred in sinusoidal PMBM machines. There are two common excitation strategies of trapezoidal PMBM drives which are called respectively as 120° and 180° electrical conduction modes. Excitation waveforms for three-phase trapezoidal PMBM have the form of quasisquare-wave (six-step) with two 60° electrical intervals of zero- current excitation per cycle for 120° conduction mode of operation. In contrast to 120° electrical conduction mode, there is no zero-current excitation period in 180° electrical conduction mode of operation. The nature of excitation waveforms for trapezoidal PMBM drives give rise to have some important system simplifications compared to sinusoidal PMBM drives. In particular, the resolution requirements for the rotor position sensor are much lower for trapezoidal machines since only six commutation instants per electrical cycle must be sensed. In addition, it has been stated in the literature that, the trapezoidal PMBM drive requires a single current sensor in the inverter dc link, but this is not the case for high power and variable-load drives [6]. Unfortunately, these simplifications leave the trapezoidal PMBM drives to face with some complex mechanisms of pulsating torque generation which don't effect their sinusoidal counterparts. In order to complete the general statement of the pulsating torque problem, it will be very convenient to give pulsating torque definitions in PMBM drive system. Any source of divergence from ideal conditions which are given before in either the motor Willor associated power converter in a PMBM drive typically gives rise to undesired torque pulsations. However, there are various specific sources for these harmonic torque components which can be defined as follows;. Cogging Torque-It is a pulsating torque component generated by the interaction of the rotor magnetic flux and angular variations in the stator magnetic reluctance. By definition, no stator excitation is involved in cogging torque production. »Ripple Torque-It is a pulsating torque component generated by the interaction of stator current magnetomotive forces and rotor electromagnetic properties which can be defined as follows: 1. Mutual or alignment torque-It is generated from the interaction of the current magnetomotive forces with the rotor magnet flux distribution. This is the dominant torque production mechanism in PMBM drive systems. 2. Reluctance torque-It is generated from the interaction of the current magnetomotive forces with the angular variation in the rotor magnetic reluctance. Surface-mounted magnet PMBM generates almost no reluctance torque..Pulsating Torque-It is the sum of cogging and ripple torque components It is clear from the definitions given above that, torque pulsation problem due to the torque production mechanism in PMBM drive systems is one of the most important research topics for that kind of drive systems. For that reason a wide variety of techniques have been proposed in literature during the past fifteen years for minimizing the generation of pulsating torque components. In chapter two of this thesis, these techniques are fully examined, the comments and results are given in details. As a summary it could be stated that these techniques can be classified in two major categories. The first major class consists of techniques related with motor design. It has no importance whether the machine is trapezoidal or sinusoidal. Basically, these techniques tend to eliminate the fundamental electromagnetic sources of the pulsating torque and optimise the design in such a way that to force it toward the ideal conditions. These motor-based techniques are reviewed in first section of Chapter two of this thesis. The second major class of techniques for minimizing pulsating torque are based on active control schemes which modify the excitation to correct for any of the non-ideal characteristics of the machine or its associated power inverter. Many of these techniques involve active elimination techniques of the pulsating torque components which would be generated using classic sinusoidal or square wave current excitation waveforms. The effectiveness of these techniques require preknowledge of the individual machine's design parameters or the use of self-tuning mechanisms to adapt to the torque production characteristics of the PMBM drive system. These approaches are basically depend on observer and estimation techniques. These controller-based approaches are reviewed in second section of Chapter two of this thesis. In this study, the target application is an outer rotor permanent magnet brushless motor which is designed and realised for a special direct drive application. As a firststage, sinusoidal-fed PMBM drives are taken into consideration. This technique is examined by using an analytical approach. It is proved that, sinusoidal excitaion of PMBM has a very complex control algorithm, and requires a precise rotor position transducer and reduces the torque value for the same frame size. As a result, the cost of the drive will be much higher than its trapezoidal counterparts. As a second step of the thesis, trapezoidal drives are folly analysed by the help of a digital simulation technique. A state space model is developed to estimate torque- speed performance of a three phase full-bridge, surface-magnet trapezoidal PMBM drive system for 120° and 180° conduction modes. The power and control electronic circuit deliver square waveforms of current. The power converter topologies of 120° and 180° conduction mode of operation are the same. While two phases are energised at any rotor position in 120° conduction mode, all of three phases are simultaneously ON in 1 80° conduction of mode operation. High frequency pulse width modulation (PWM) of the lower bridge transistors is used to control speed and torque. Broadly speaking, the state space model includes simulations of rectifier/filter or DC link, the resistance/self/mutual inductances^ack EMF circuits of the motor and switching patterns of transistor bridge. They interact at the DC link filter capacitor and their governing equations are solved numerically by using a fourth order Runge- Kutta algorithm in which current or torque is the dependent variable. Thus not only the behaviour of the inverter but also the behaviour of the rectifier are taken into consideration to find out the influences of the drive on torque pulsation. As a result, the generalised set of differential equations which covers PWM technique are obtained and analysis of the overall drive system is described in principle. Analysing the previous studies on pulsating torque minimization in PMBM drives and results of digital simulation of six-step trapezoidal techniques, a new approach is proposed which is called twelve-step excitation of trapezoidal PMBM drive system. The only modification needed is three additional Hall-effect sensors. The new switching scheme works with almost any trapezoidal brushless motor, regardless of the number of poles, phases or motor design configuration. The advantage of twelve-step excitation lies in the increased number of MMF vectors produced during electronic commutation. The six additional vectors reduce torque pulsation when two fields are in quadrature. Unfortunately, adding six more vectors is not the complete solution. The amplitude of torque vectors produced by two conducting phases are not the same with those generated with three phase conduction mode. It could be stated that, if all twelve vectors are not made equal in amplitude, the additional torque vectors can produce a higher frequency torque ripple. For example, in conventional six step excitation, two phases are energised at any time instant and the resultant field is midway between related phases with a magnitude of 1.73 pu. Similarly for twelve-step excitation, if we take one high and two low leg switches conduction, the phase which is connected to dc bus will produce current for the other phases connected to power ground. The amplitude of the resultant torque vector is 1.5 pu in that case. As a result, the unbalanced vectors generate the torque pulsation but can be eliminated by making all vectors produce the same torque vector using current control. In twelve-step excitation system, it could be compensated duringthree phase ON stages, with phase current which is amplified by a factor of 1.153 to make all vectors equal in amplitude. In order to investigate the behaviour of the new approach, the same procedure which is used for six step techniques, is applied. In other words, the same mathematical modelling approach is also developed for twelve-step excitation of trapezoidal PMBM drives. The simulation results are compared with those of six-step 120° and 180° electrical conduction modes. To have a criteria for comparison, the pulsating factor of torque is defined as: T -T M= j Xl0° 0) The results are very promising and the pulsating factor of torque is decreased by a factor of 50% especially in low speed range. Trapezoidal drive prototypes with six and twelve-step techniques are also designed and realised by using a 8 bit microcontroller-PLD (programmable logic device) based electronic circuits. The average torque values are measured on a dynamometer to compare with those of simulations results. A good agreement between the experimental and analytical results has been observed except very low speeds. In addition to a new excitation approach which is called as "Twelve-Step Trapezoidal-Drive of Permanent Magnet Brushless Machines", a digital current control technique is also introduced to have a reliable drive with special features dedicated to twelve-step excitation approach. It can also be used for 120° conduction mode of six-step trapezoidal drive. A digital technique of current regulation is preferred to ensure a current demand which is the upper limit of hysteresis controller. In order to define the analytical equations of the phase current, the idealised linear behaviour of the hysteresis controller is taken into consideration. The analytical equations of the phase current is proposed in details for all twelve stages of excitation. The combinations of motor phases energised due to the rotor position, is modelled as an equivalent inductance in series with an equivalent back EMF function. Assuming that the DC link voltage exceeds the back-EMF the phase current follows a linear trajectory according to equations which are introduced for every discrete excitation steps. Note that the same technique can also be applied to six-step excitation approaches with different converter topology and equivalent back EMF and inductance value. By combining the advantages of twelve-step excitation, this current control algorithm provide the following key features;. Elimination of all discrete current sensors except one and single one located on dc link. Protection against high circulating current loops during commutation »High degree of drive circuit integration, minimizing the number of drive electronics components. PWM frequency is constant In conclusion, the smooth torque production in trapezoidal PMBM drives is achieved by a package solution which is a combination of a different excitation and current- control strategy.

Abstract

Tez (Yüksek Lisans) -- İstanbul Teknik Üniversitesi, Fen Bilimleri Enstitüsü, 2014Thesis (M.Sc.) -- İstanbul Technical University, Institute of Science and Technology, 2014Teknolojinin gelişmesiyle birlikte kullanım alanları da artan elektroniğin getirdiği yenilikler günlük hayat ile sınırlı kalmayıp endüstriyel sistemleri de kapsamıştır. Servo sistem de bunlardan biridir. Bu tez kapsamında sabit mıknatıslı senkron motorların servo uygulamalarında PID kontrolör parametrelerinin otomatik ayarlanması işlenecektir. Servo sistem genel olarak motor sürücü, kontrolör ve servo mekaniği olmak üzere üç başlık altında incelenebilir. Motor sürücü kısmını ele alırsak, bu kısım şu birimlerden oluşur: elektrik motoru, elektronik komütatör ve sensörleri. Elektrik motoru olarak alternatif akımlı sabit mıknatıslı senkron motor kullanılmıştır, elektronik komütasyon için vektör kontrol algoritması ve PWM üretimi için de uzay vektör modülasyonu yöntemi kullanılmıştır. Üretilen PWM IPM tabanlı kuvvetlendiriciye girmiş ve kuvvetlendirici tarafından motor fazları sürülmüştür Vektör kontrol algoritması çalışmak için rotorun açısal pozisyonuna ihtiyaç duyar ve bunun ölçülmesi için 2500 artımlı mutlak pozisyonlu dördül enkoder kullanılmıştır. İkinci olarak da dijital kontrolör kısmını ele alırsak, bu kısım da mikrokontrolör devresi, haberleşme birimleri ve kontrolör yazılımından oluşur. Mikrokontrolör devresinde mikro işlemci tabanlı bir dijital sinyal işleyici (DSP) ve bunun çalışması için gerekli diğer elektronik devre elemanları bulunur. Haberleşme birimi olarak Visual C# dilinde kullanıcı arayüzü yazılmıştır. Kontrolör yazılımında i_q akımını, i_d akımını ve motor hızını kontrol etmek için 3 farklı dijital PID kontrolör vardır. Üçüncü olarak ise ve servo mekaniğinden bahsedebiliriz. Motor, elektronik devreler, sensörler ve eğer gerekliyse dişli kutusu gibi parçaları birbirlerine sabitleyen ve tüm sistemi toz, su, darbe gibi dış etkilerden koruyan kısımdır. PID hız kontrolör parametrelerinin otomatik ayarlanması için gerekli olan ilk adım sistem modelinin elde edilmesidir. Bunun için sisteme uyarı sinyali yollanır ve sistem cevabına bakılır, uyarı sinyali olarak sisteme 10 saniye boyunca değişken frekanslı sinüs sinyali uygulanmış ve bu süre zarfında sinüs sinyalinin frekansı 0,5Hz’den başlanarak lineer olarak 10Hz’e kadar artırılmıştır. Ardından ISE kriterini sayısal olarak optimize edecek şekilde sistem parametreleri bulunmuştur. Bulunan bu parametreler hız kontrolörü tasarımında kullanılmıştır. Hız kontrolörü için dijital PID kontrolör kullanılmış ve örnekleme periyodu 0,001Hz olarak belirlenmiştir. Sistem modelinin parametreleri yüzde aşım, yerleşme zamanı gibi kriterlerin de dâhil edildiği IKH maliyet fonksiyonunu minimize edecek şekilde aranmıştır. Algoritmalar farklı sistemler için çalıştırılmış, simülasyon ve gerçek test sonuçları incelendiğinde kullanılan yöntemin başarılı olduğu görülmüştür.Development of the technology also brought the improvement of the electronics and made it part of the daily life. It also entered to the industrial area and the servo system is one of the results of industrial development of electronics. Purpose of this theses is auto tuning of PID speed controller parameters for permanent magnet synchronous motor servo applications. Permanent magnet synchronous motors are kind of brushless motors and they have several advantages and disadvantages when they compared with the conventional brushed motors. Advantages of brushless motors are; no electrical sparks, voltage drops and EMI generation because of mechanical brushes, they can work in oil and dust, they can work long time without any maintenance, they produce higher torque, they have less weight and they have less size. Disadvantages of brushless motors are; need of electronic commutator and higher price. It can be easily seen that, advantages of the brushless motors are more than the disadvantages and this is the reason why usage of brushless motors in industry increases and usage of brushed motors decreases. Brushless motors have two different types one of which is brushless DC motor and the other is brushless AC motor. Brushless DC motor, also named as permanent magnet DC motor, is 3-phase motor and it has trapezoidal back EMF, this means that when it is rotating it generates voltage shaped as trapezoidal. Because of this trapezoidal back EMF it is needed to be powered with the square wave voltage and it has own driving procedure that is named as 6-step commutation. The other type of brushless motors is brushless AC motor and it is also called as permanent magnet synchronous motor. This is also 3-phase motor but difference of this type is sinusoidal back EMF. This means that when it is rotating it generates voltage shaped as sinus function. Because of this sinusoidal back EMF it is needed to be powered with the sinus wave voltage and it has own driving procedure that is named as vector control. When compared with the brushless DC motor, permanent magnet synchronous motors have some important advantages over them. Because of the sinusoidal back EMF, they produce less EMI and their torque output has fewer ripples. Usage of the vector control algorithm also increases the dynamic load performance and this is the reason why washing machine producers use permanent synchronous motors, this motor kind gives the best performance for these kinds of nonlinear loads. Permanent synchronous motors are synchronous motors and that means that magnetic flux produced with the rotor magnets and the stator windings must be at the same frequency. They also need to be orthogonal (90º) to each other for the maximum torque production. Servo system that is designed in this project is a digital servo system, this means that its controller is not analog controller, instead it has microprocessor based digital controller. In general, digital servo system can be divided in three main parts, first part is motor driver, second part is controller and the last part is servo mechanic. First part is the motor driver part and it has sub parts as permanent magnet synchronous motor, electronic commutator and amplifier. Electronic commutator and amplifier take the function of brushes in brushless motors. It calculates the commutation frequency, phase and voltage amplitude, then it generates PWM signals and amplifier amplifies these signals to power the motor phases. Motor used in this thesis is 220V motor and nominal RMP is 5000. It has integrated absolute quadrature encoder with 2500 increment, this makes the resolution of 10000 increments. Encoder is the part of commutation electronics because commutation algorithm needs to know the actual motor angular position in every step. Commutation procedure used in this thesis called as vector control and also named as field oriented control (FOC) algorithm. This algorithm can be summarized in seven steps. First step is measurement of the phase currents, only two of three phase currents is sufficient because third one can be found with the usage of the other two. Second step is Clarke Transform, that converts three phase currents to a two-axis system. This conversion produces i_α and i_β from i_a, i_b and i_c. Third step is Park Transform, that converts two axis orthogonal i_α and i_β system to the new two axis orthogonal system that rotates with the rotor flux. This conversion produces i_d and i_q currents. I_d current produces magnetic flux and independent from torque production, i_q current produces torque and independent from flux production, this allows controlling the produced flux and torque independently Fourth step is PID control loop. Control loop includes two different PID controllers, first one controls the i_d current and the second one controls the i_q current, purpose is maket the i_d current zero and track the refference i_q command. I_q controller produces V_q reference as an output and i_d controller produces V_d. Fifth step is finding the θ (angular position of the rotor), there are two different methods to find θ. First one is sensorless algorithm and it only uses V_α, V_β, i_α and i_β for estimation. Second method is using absolute positon sensor and reading sensor data, this is also the chosen method for this project because it gives more precision results at the full range of speed. Sixth step is Inverse Park Transform, that transforms V_q and V_d back to the stationary reference frame using the new θ value, that finds the V_α and V_β commands. Seventh and the last step is converting V_α and V_β commands to the motor phase voltages V_a, V_b and V_c. This is done with the space vector algorithm and it produces the PWM duty cycles to produce the destination voltages. Second part is controller part and it has also some sub parts as, digital controller board, user interface and digital controller code. Digital controller board designed for this thesis and it includes 32-bit, 150MHz, floating point Texas Instruments C2000 family Delphino series TMS320F28335 digital signal processor (DSP). This board also includes ASRAM, EEPROM, linear and switchmod regulators, analog and digital buffers, two CAN, one RS232 and one RS485 interface, two quadrature encoder interface, analog inputs for ADC, external digital inputs and outputs. Board is six layer PCB and gold plated, all connector pins have ESD and EMI protection filters. The other sub part of controller is user interface, it is a computer program and used for testing the system and setting some EEPROM parameters. It is written in Visual C# language and communicates with servo system via RS232 port. More important sub part of controller section is digital controller part, that includes control algorithm and other DSP codes. It has three control loops, first one is speed control, second one is i_q current control and third one is i_d current control. All controllers are digital PID controllers. Speed controller works at the 1 kHz and two current controllers works at the frequency of 1 kHz. Speed controller gives its output to the i_q controller as the reference value, i_d controller works for ragulate the i_d at the zero value. Servo system mechanic is not explained in detailed in this thesis because it is out of the scope of this thesis. It holds all the mechanics and electronic parts of the servo system together and prevents them from dust and damage. Last part of this thesis about the auto tuning of the PID speed controller. To control the system, firstly system model needs to be known. Because of the change of the system model with the usage area of the servo system, system model is also needed to found automatically. To find the system model, controller produces excitation i_q signal. Most common used two types of excitation signals are multi-sine and chirp signals. Chirp signal is used in this thesis and its frequency changes between 0,5Hz – 10Hz in 10 seconds, its amplitude is 0,05 which is chosen for not to damage the system with high power vibrations. Numerical method used to find the best fit transfer function parameters and ISE cost function is used for this purpose. System transfer function is first order and so finding the two parameters is sufficient to modeling the system. One of these two parameters (K) are searched at 201 points and the other one (A) is searched at 401 points because it is logarithmic parameter and precision is more important from the other linear parameter that searched at 201 points. Starting and ending points of searching points are found according to the general 750W motor parameters, this means that parameters searched around the parameters of the common motor model parameters. But search area is too wide not to be affected from large parameter changes. Then these system parameters are used to find PID controller parameters. PID speed controller is found with the usage of cost function like in optimal control. In addition to the ISE criterion some additional rules like overshoot and settling time is used. To protect the PID controller from wind up, anti-wind up mechanism is used. Controller output is saturated to be similar with the real system. Motor model estimation algorithm and PID auto tuning algorithm is tested via Matlab m-function and results are simulated at the Matlab/Simulink pocket program. Results show that both of model estimation and PID auto tuning algorithms are successful. For the system that has gain of 84 and time constant of 0,1 real K and A parameters of the discrete time model are 0,8358 and 0,99 respectively. Estimated model parameters found as 0,8312 for K and 0,99 for A, step responses of two system is nearly identical. For the worst result, gain error found with the %3,75 error and this is also negligible. When auto tuned PID controller’s performance is analyzed, it is seen that first controller gives 0,0015 second settling time for %2 band, no steady state error and nearly no overshoot. Second controller is designed for soften the system response and it gives %1,4 overshoot, 0,045 second settling time and no steady state error.Yüksek LisansM.Sc

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